High probability of intercept radar detector

ABSTRACT

Detecting continuous wave police radar includes receiving an input signal from a first antenna, the input signal comprising a continuous wave emission within at least one radar band; sweeping a composite local oscillator signal through a range of frequencies from a first frequency to a second frequency in a predetermined time period so that the composite local oscillator signal has a first chirp rate with a first chirp rate magnitude of between 0.15 MHz/μs and 3.5 MHz/μs or even higher; and mixing the input signal from the first antenna with the sweeping composite local oscillator signal to produce an output signal having an intermediate frequency. A next step can include determining that the input signal from the first antenna includes a police radar signal based on the output signal.

RELATED APPLICATIONS

The present application is a continuation of, and claims benefit from,U.S. application Ser. No. 13/834,905, filed Mar. 15, 2013, entitled HIGHPROBABILITY OF INTERCEPT RADAR DETECTOR, the contents of which areincorporated herein by reference in their entirety.

FIELD OF THE INVENTION

The present invention relates generally to police radar detectors usedin motor vehicles and, more particularly, to decreasing detector sweeptime when detecting radar signals.

BACKGROUND

Conventional radar detectors almost universally employ scanningsuper-heterodyne receiver architectures. To achieve good sensitivitywith lower cost, conventional radar detectors tend to sweep relativelyslowly, often requiring several tenths of a second to sweep a coveredspectrum. As a result, some radar gun manufacturers have developed radargun designs that transmit very brief pulses as a technique for avoidingdetection. The brief transmission is of a relatively short duration andmay be conveniently referred to as a “POP transmission” which is aphrase coined and trademarked by MPH Industries. A detector with sweepperiods lasting several tenths of a second is likely to entirely miss aradar gun transmission that lasts only in the neighborhood of no morethan several tens of milliseconds. One approach for a detector designmay involve sweeping the spectrum much faster to try to intercept thesebrief transmissions. In this way, the detector will tune through thetransmission frequency during the interval that the radar signal isactually present. However, this approach will greatly increase therequired bandwidth of the detector's receiver. Because the receivedsignal power remains unchanged in the increased bandwidth, thesignal-to-noise ratio degrades and a commensurate decrease in thresholdsensitivity occurs.

Thus, there remains a need for another approach to reliably detectingthese brief radar gun transmissions that avoids undue loss of thesignal-to-noise ratio.

SUMMARY

One aspect of the present invention relates to a detector for detectingcontinuous wave police radar that includes a first antenna configured toreceive an input signal, the input signal comprising a continuous waveemission within at least one radar band, and a composite localoscillator configured to sweep a signal through a range of frequenciesfrom a first frequency to a second frequency in a predetermined timeperiod to produce a composite local oscillator signal having a firstchirp rate with a first chirp rate magnitude of at least 0.15 MHz/μs.The detector also includes a mixer configured to mix the input signalfrom the first antenna with the sweeping composite local oscillatorsignal to produce an output signal having an intermediate frequency; anda signal analyzer is configured to determine whether the input signalfrom the first antenna includes a police radar signal based on theoutput signal.

Still a further aspect of the present invention relates to a method ofdetecting continuous wave police radar that includes receiving an inputsignal from a first antenna, the input signal comprising a continuouswave emission within at least one radar band; sweeping a composite localoscillator signal through a range of frequencies from a first frequencyto a second frequency in a predetermined time period so that thecomposite local oscillator signal has a first chirp rate with a firstchirp rate magnitude of at least 0.15 MHz/μs; and mixing the inputsignal from the first antenna with the sweeping composite localoscillator signal to produce an output signal having an intermediatefrequency. A next step in the method can include determining that theinput signal from the first antenna includes a police radar signal basedon the output signal.

BRIEF DESCRIPTION OF THE DRAWINGS

While the specification concludes with claims particularly pointing outand distinctly claiming the present invention, it is believed that thepresent invention will be better understood from the followingdescription in conjunction with the accompanying Figures, in which likereference numerals identify like elements, and wherein:

FIG. 1A depicts a block-level diagram of a radar search receiver withchirp compression.

FIG. 1B depicts a block-level diagram of portions of a radar signaldetector in accordance with the principles of the present invention.

FIG. 2A depicts major components of the Ka-band signal path from thefront antenna of the radar signal detector of FIG. 1B.

FIG. 2B depicts major components of the K-band signal path from thefront antenna of the radar signal detector of FIG. 1B.

FIG. 2C depicts major components of the Ku-band signal path from thefront antenna of the radar signal detector of FIG. 1B.

FIG. 2D depicts major components of the X-band signal path from thefront antenna of the radar signal detector of FIG. 1B.

FIG. 3A depicts an exemplary continuous wave radar signal such as thoseemitted from a police radar gun.

FIG. 3B and FIG. 3C depict, respectively, a linear down-chirp signal anda linear up-chirp signal in accordance with the principles of thepresent invention.

FIG. 3D depicts a compressed pulse resulting from propagating a linearchirp signal through a matched filter in accordance with the principlesof the present invention.

FIG. 4A depicts a conceptual graph of an up-chirp.

FIG. 4B depicts a conceptual graph of a down-chirp.

FIG. 4C depicts a graph of the frequency-to-time ratio of the sweptlocal oscillator when sweeping through the KA-band in accordance withthe principles of the present invention.

FIG. 5 depicts the general structure of the local oscillator waveformfor a particular radar band sweep in accordance with the principles ofthe present invention.

FIG. 6 depicts a flowchart of an exemplary sweep process in accordancewith the principles of the present invention.

FIG. 7 depicts representative signal shapes for a detected RF signal andan undesired image signal.

FIG. 8A depicts a flowchart of an exemplary process to implementing afixed detection window in accordance with the principles of the presentinvention.

FIG. 8B depicts a block level diagram of utilizing a dynamicallyadjusted noise floor in accordance with the principles of the presentinvention.

FIG. 9 depicts what the activation of a radar gun may look like whendetected in accordance with the principles of the present invention.

DETAILED DESCRIPTION

In the following detailed description of the preferred embodiment,reference is made to the accompanying drawings that form a part hereof,and in which is shown by way of illustration, and not by way oflimitation, specific embodiments in which the invention may bepracticed. It is to be understood that other embodiments may be utilizedand that changes may be made without departing from the spirit and scopeof the present invention.

An inherent property of a matched filter is that it has an impulseresponse that is a time-reversed replica of the desired signal (plus atime delay required for causality). In a wideband search applicationusing a compressive receiver design similar to one as described below,one example desired signal structure is a linear chirp. Compressivereceiver design may reasonably begin with selection of a compressivefilter since it is one of the most challenging receiver components. Oneexample compressive filter type is a dispersive delay line (DDL) surfaceacoustic wave (SAW) filter which is generally fabricated to have animpulse response that is a down-chirp rather than an up-chirp becausethe down-chirp devices tend to deliver better performance (e.g. lowerloss) than do up-chirp devices. Assuming selection of a DDL SAW filterhaving the favored down-chirp impulse response, the matched-signal inputwould be an up-chirp and accordingly the receiver design would includean oscillator swept so as to deliver up-chirps of the proper rate to aninput of the DDL SAW filter.

An example super-heterodyne receiver architecture described belowlinearly sweeps the local oscillator (LO) to down-convert receivedcontinuous wave radio frequency (CW RF) signals into linear up-chirps atthe receiver's intermediate frequency (IF) output (i.e. the matchedfilter input). In one example embodiment described herein, thereceiver's RF frequency conversion conforms to the formula: IF=RF−N*LOwhere N=2, 3, 5, 7 to sweep X, Ku, K, and Ka, respectively. Therefore,to deliver signal up-chirps at the IF, the LO generates down-chirps. Oneof ordinary skill will recognize opposite behavior can be designed andaccomplished if the compressive filter conversely has an input responsethat happens to be an up-chirp.

By way of example, a receiver design may include a DDL filter that has amatched signal input chirp rate of +3.5×10¹² Hz/s, with center frequencyof 280 MHz. In accordance with matched-filter theory, the filtersimpulse response is a burst down-chirp centered at 280 MHz, with a chirprate of −3.5×10¹² Hz/s. The duration of the impulse-response burst isestablished by specifying the required dispersive time-delay length ofthe filter. This dispersive delay, or dispersive time-delay length, isthe differential delay between the travel times of the frequencyextremes of the DDL filter. The acoustic velocity on a surface of a SAWis essentially constant for all frequencies; however, the effective pathlength to an array of resonators metalized on the surface varies withfrequency. Thus, constant velocity over varying lengths gives rise tovarying delay times for different frequencies. Subsequent descriptionrefers to “delay” for convenience, but it should be understood thatdispersive time-delay length is the salient parameter under discussion.The bandwidth of the impulse burst is simply the dispersive time-delaylength times the chirp rate and this bandwidth can also be characterizedas the signal bandwidth of the filter. The dispersive time-delay lengthis one performance parameter of the filter that governs the amount ofsignal energy that can be captured by the filter and, thus, governs thefilter's performance advantage. In principle, a receiver's performanceimproves with an increase in the time-delay length of the filter. Forexample, a filter having a dispersive time-delay length between about 2μs to about 20 μs provides beneficial results in the frequency rangestypically encountered with police-band radar signals. In particular, anexample dispersive time-delay length of 4 μs is used in an exampledesign described below to help facilitate understanding of theprinciples of the present invention.

FIG. 1A depicts a block-level diagram of a radar search receiver withchirp compression. For clarity, intermediate components such as filtersand amplifiers are omitted from the idealized diagram of FIG. 1A. Acontinuous wave radar signal is received by an antenna 52 and is fed, assignal “RF” to a mixer 54. An example of such a signal is depicted inFIG. 3A which shows a fixed frequency signal pulse 300 over a period oftime.

Another input to the mixer 54 is a swept local oscillator signal or “LO”signal generated by a linear sweep generator 60. The linear sweepgenerator 60 generates a signal whose frequency changes in a linearfashion over time. If the frequency of the generated signal increases astime progresses, then this is conveniently referred to as an “up-chirp”.If the generated signal decreases in frequency as time progresses, thenthis is conveniently referred to as a “down-chirp”. As used herein, adown-chirp signal may be described as having a chirp rate of X Hz/sbecause the term “down-chirp” provides an indication that the chirp ratefrequency is decreasing at a rate having a magnitude of X Hz/s.Equivalently, for clarity, a down-chirp signal, even when identified asa “down-chirp”, can be described as having a chirp rate of −X Hz/s as away to emphasize its decreasing rate. FIG. 3B depicts a down-chirp sweepsignal 302 that may be produced by the linear sweep generator 60 andFIG. 3C conversely depicts an example up-chirp sweep signal 303. FIGS.4A and 4B depict respective chirp signals as a graph of frequency versustime. The signal of FIG. 4A is an up-chirp signal with a positive chirprate 402 that produces an increasing frequency as time progresses; whilethe signal of FIG. 4B is a down-chirp signal with a negative chirp rate404 that produces a decreasing frequency as time progresses.

When the RF signal and the LO signal are combined at the mixer 54, thenthe RF signal is frequency shifted by the instantaneous frequency of theLO signal. In particular, mixing an RF signal having a frequency f_(RF)and a LO signal having a frequency f_(LO) produces two different signalsat a respective intermediate frequency. One of the produced signals willhave an intermediate frequency of f_(RF)+f_(LO) and the other producedsignal will have an intermediate frequency of f_(RF)−f_(LO). In FIG. 1A,the “minus” sign next to mixer's input for the LO signal indicates thatthe filter of FIG. 1A happens to pass f_(IF)=f_(RF)−f_(LO). It shouldalso be noted that f_(IF) is proportional to −f_(LO); as a result, adown-chirping LO will be reflected as an up-chirping signal in the IFchirp output.

When a continuous wave signal such as that in FIG. 3A is mixed with adown chirp signal as shown in FIG. 3B, the resulting signal, “IF CHIRP”,from the mixer 54 will be an up-chirp signal similar to the signal 303of FIG. 3C. A filter 56 is selected that is a matched filter withrespect to the shape of the IF CHIRP signal. As described earlier, amatched filter is one that has an impulse response with exactly the sameshape as its input signal but reversed in time with an added delayrelated to the duration of the input signal.

Chirp signals are characterized by a chirp rate, as shown in FIGS. 4Aand 4B. For example, the linear sweep generator 60 may produce a signalwhose frequency linearly changes at a rate of −3.5 MHz/μs. Equivalently,this signal would have a chirp rate of −3.5×10¹² Hz/s. The invertingarrangement of the mixer 54 results in the signal at the IF output(i.e., the desired signal) having a chirp rate of 3.5×10¹² Hz/s. As formatched filter selection, the matched filter impulse response for achirp signal is itself a chirp signal having the same magnitude but withan opposite sign than that of the desired signal. Accordingly, for theexample signal just described, the matched filter would have an impulseresponse chirp rate of −3.5×10¹² Hz/s; alternately stated, the filter'smatched signal is an up-chirp at 3.5×10¹² Hz/s, precisely what isdelivered by the sweep generator/mixer combination. This particularchirp rate is provided merely by way of example and other chirp rateshaving different magnitudes, either greater than or less than 3.5×10¹²Hz/s, are contemplated as well.

The result of the up-chirp linear FM chirp signal 303 propagatingthrough the matched chirp compression filter 56 will be a compressedpulse similar in shape to the signal 304 of FIG. 3D. In particular, thesignal 304 is a sinc waveform having a peak occurring at a particulartime T 306 that is based on the frequency of the RF signal. Accordingly,a detector 58 can detect an occurrence of a peak in the signal 304 andconvey this occurrence to control and analyzer circuitry 62. The controland analyzer circuitry 62 can then determine if the detected signal islikely a received radar transmission and also determine the frequency ofthat radar transmission. The control and analyzer circuitry 62 can alsobe coupled with the linear sweep generator 60 to control the LO waveformthat is being provided to the mixer 54.

Additionally, a user interface 63 can include audible or visualindicators that are representative of the signals being received anddetected as well as allow a user to select different operatingcharacteristics of the device of FIG. 1A.

Applying these principles to the detection of radar gun transmissions,there is a continuous wave signal at an unknown frequency somewhere inthe radar bands, a spectrum that can total about 3 GHz. Asuper-heterodyne receiver remains a viable technique for scanningthrough this wide spectrum. As a result, the received RF signal will bedown-converted to an intermediate frequency signal that is a linearchirp. Thus, a matched filter will also have an impulse response that isa linear chirp. Such a matched filter can be realized as a dispersivedelay line acting as a receiving filter. In this configuration of areceiving filter, as the intermediate frequency chirp propagates throughthe dispersive delay line, the delay line accumulates the energy in thematching chirp during the several microseconds the signal is presentwithin the delay line. The delay line acts to compress the matched chirpand delivers the stored energy as a narrow output pulse only a few tensof nanoseconds wide.

A matched filter which captures as much signal energy as possibleprovides the best sensitivity. However, in practice, a radar detector isdesired that will intercept very brief radar pulses and, thus, a sweepchirp rate should be fast enough to cover the entire spectrum in lessthan the length of a POP transmission. The length of a POP transmissionmay, for example, be about 16 ms. In this example, the entire sweepcycle should be completed in less than 16 ms to ensure interception ofthe limited duration POP transmission. This principle should governirrespectively of the number of antennae employed or the number of radarbands swept. To elaborate, for reasons of economy, it may be desirableto share the DDL filter and other receiver circuitry sequentiallyamongst multiple bands and/or antennae. Receiver sensitivity might haveto be compromised commensurately, but completing the sweep cycle oftenenough to ensure POP signal interception should be the guidingconstraint. These considerations will be evident to those havingordinary skill. Some circumstances may motivate sweeping faster than 16ms. If both a front and rear antenna are considered, in a sequentialmanner, then the sweep should be able to be performed in less than 8 ms.If multiple sweeps are desired in order to identify spurious signals orfalse alarms, then even faster sweep rates are beneficial. FIG. 1Bdepicts a block-level diagram of portions of a radar signal detector inaccordance with the principles of the present invention. There are threeparticular radar bands that it is typically desirable to sweep throughin order to detect radar gun transmissions. The X-band center isnominally at 10.525 GHz but a desirable sweep will include the range of10.49-10.56 GHz; The K-band center is nominally at 24.150 GHz but adesirable sweep will include the range of 24.035-24.265 GHz; and theKa-band center is nominally at 34.7 GHz but a desirable sweep willinclude the range of 33.35-36.08 GHz. A fourth band may also be swept:the Ku-band center is nominally at 13.45 GHz but a desirable sweep willinclude the range of 13.38-13.52 GHz. These frequencies encompass theFCC allocated police radar bands plus modest “over-scan” for safety.

The diagram of FIG. 1B shows different switches (e.g., 112A, 112B, 118,126, 158) that define the signal path through the radar signal detectorat a particular time. These switches may be implemented in a variety ofdifferent ways without departing from the scope of the presentinvention. For example, one or more of the switches may be a transistorhaving its gate voltage set so as to be transmissive or non-transmissivein a given direction or various switches can be functionally achieved byactivating a bias on amplifiers in a desired path and deactivatingamplifiers in unwanted paths. The microprocessor 180 and/or fieldprogrammable gate array (FPGA) 176, or similar control circuitry, cancontrol the switches so as to select a current signal path and, thereby,select a particular band of radar signals of interest at a particulartime.

When considering signals arriving at a front antenna 102, the switches112A and 112B will select the appropriate signal path. Thus, an RFsignal at the front antenna 102 will pass to the diplexer 104 and besplit into K-band and Ka-band frequencies on one path and X-band andKu-band frequencies on a different path. The K-band and Ka-bandfrequencies can be amplified by an amplifier 106 and mixed with a localoscillator based signal at a mixer M1F 108. The output of the mixer 108can be amplified by amplifiers 110, 116, and 128 before propagating toanother diplexer 144. At the diplexer 144, the K-band signals areseparated from the Ka-band signals. K-band signals pass through theswitch 158 and are amplified by an amplifier 160 before being mixed, ata mixer M4 162, with a local oscillator signal from the voltagecontrolled oscillator 140. The output F_(O) 164 of the mixer M4 162 isamplified by an amplifier 168 and propagated to a dispersive delay linefilter 170. One of ordinary skill will recognize that the differentbands can be combined and separated in ways that are different thanthose of FIG. 1B. For example, the K-band signals could be handled by aseparate signal path while the Ka and X-band signals can be in acombined path from which they are alternately selectable.

At the diplexer 144, the desired band of signals could have been in theKa-band which would have resulted in Ka-band signals being propagatedfrom the diplexer 144 to a band-pass filter 146 and mixed, at a mixer M3148, with a local oscillator based signal. The output from the mixer M3148 can be filtered with another band-pass filter 150 and can then passthrough the switch 158 to the amplifier 160 where it is then mixed inthe mixer M4 162 with a local oscillator signal from the voltagecontrolled oscillator 140. The output F_(O) 164 from the mixer M4 162can then be amplified using the amplifier 168 and propagated to thefilter 170. At the mixer M4 162, an image signal 166 can be rejected by,for example, being shunted to ground or other known techniques.

Returning to diplexer 104, the other path is for received RF signals ineither the X-band or the Ku-band. Signals in either of these bands maybe amplified by an amplifier 107 and then a switch 118 can directKu-band signals directly to an amplifier 124 or can direct X-bandsignals to an amplifier 120. The output of the amplifier 120 is mixed atmixer M2 122 with the local oscillator signal from the voltagecontrolled oscillator 140 and propagated to the amplifier 124. A switch126 can connect the output from the amplifier 124 to the amplifier 128so that the diplexer 144 can direct the X-band and Ku-band signals tothe band-pass filter 146 and from there to the mixer M3 148. At themixer M3 148, the output from the filter 146 and a local oscillatorbased signal are mixed and propagated to the band-pass filter 150. Theoutput from the filter 150 passes through the switch 158, the amplifier160, the mixer M4 162, and the amplifier 168 to arrive at the filter170.

When considering signals received by a rear antenna 138, similar signalpaths are provided for each of the different bands of signals in thecovered spectrum. However, these alternative signals paths utilize adiplexer 136, various amplifiers 132, 134, and 114, and a mixer M1R 130.

The circuit arrangement of FIG. 1B utilizes a single voltage controlledoscillator 140 to produce a local oscillator signal F_(LO). As is knownin the art, a reference voltage generator 156, phase lock loop 152, andamplifier 154 work in conjunction with the voltage-controlled oscillator140 to produce a linear sweep signal having desired characteristics forthe band of signals being detected at a given moment. The signal F_(LO)may be doubled by a frequency doubler 142 and provided to various onesof the mixers (e.g., M1F, M1 R, M3).

The microprocessor 180 and/or the FPGA 176 coordinates frequency controland receiver band switching. In particular, as mentioned, the frequencyof the VCO 140 is controlled by the PLL 152 and reference generator 156.An example reference generator 156 may be a direct digital synthesizer(DDS) that generates a dynamically varying reference input to the PLL152. The PLL 152, in turn, is used to frequency multiply the referenceinput to an appropriate frequency for F_(LO). Example devices may beAnalog Devices AD9913 and ADF4106 for the DDS and PLL, respectively.Alternatively, Analog Devices ADF4158 may be used as the referencegenerator 156 that is capable of synthesizing sweep waveforms.

Ultimately, a received signal from any of the bands and from either thefront or rear antenna will be propagated to the dispersive delay linefilter 170. This filter can be implemented using a surface acoustic wavedevice or by digitizing the IF with an analog-to-digital converter andprocessing with an appropriately programmed digital signal processor. Aswill be discussed in more detail below, the filter 170 is a filter thatis matched with the shape of the signal F_(O) produced as describedabove. The filter 170 produces an output that is detected by ademodulating log amplifier 172. The output of the log amplifier 172 iscontinuously digitized by a high-speed analog-to-digital converter 174and input to the FPGA 176. The FPGA 176 digitally processes this inputand determines if the output from the converter 174 exceeds a noisefloor sufficiently to be deemed a real, detected signal.

An algorithm within the FPGA 176 may dynamically quantify the noisefloor so that a detection threshold can adapt to fluctuations in noisepower as the bands are being swept. When the FPGA 176 recognizes asignal, it stores (for example, in a FIFO buffer) the signal's peakamplitude as a measure of signal strength and attaches a timing “tag”that indicates the time when the peak was detected. If a side lobe 308is falsely considered to be a detected pulse, then the pulse waveform304 of FIG. 3D may inaccurately be detected as multiple closely-spacedsignals. Accordingly, the FPGA 176 may use a detection window approachto avoid inaccurate results. The detection window is a period of timeduring which the FPGA 176 monitors the output from the converter 174 tosearch for the peak amplitude. In one embodiment, the FPGA 176dynamically analyzes the converter output to determine when a pulse isover. A simpler approach is to have a fixed-size detection windowwherein the window size is based on the expected pulse width at largesignal strengths and a desired resolution of nearby signals.

The detector of FIG. 1B may also include a laser detector 184. Inparticular, the laser detector 184 can include a front sensor and a rearsensor for detecting police LIDAR signals. A power supply 182 can beincluded to ensure sufficient electrical power for all the componentsand a microprocessor 180 can be included to control the operation andtiming of the appropriate elements of the detector of FIG. 1B.

In FIG. 1B, the radar signal detector was depicted in such a way as tohighlight how a single local oscillator 140 is used to produce all ofthe mixer signals for all of the radar bands in the covered spectrum.This scheme offers a number of advantages with respect to hardwareefficiency. Especially advantageous is the avoidance of multiple fixedLOs and their propensity to generate problematic mixing products thatresult in internal spurious signals. However, it should be obvious thatnumerous other receiver frequency schemes could be adopted withoutdeparting from the spirit of the invention.

To explain operation of the circuitry in FIG. 1B, FIGS. 2A-2D areprovided below to isolate each signal path for a respective band ofradar signals. In these figures, the various switches, amplifiers, andfilters have been omitted for clarity. For each radar band, the VCO 140produces an appropriately swept F_(LO) that is different from the F_(LO)for the other bands. As depicted in FIG. 2A, a particular spot F_(LO)frequency (e.g., 5.1029 GHz) is shown corresponding to a particular RFsignal frequency (e.g., 36 GHz) within a particular radar band (e.g.,Ka-band).

Within each of the signal paths depicted in FIGS. 2A-2D there is arespective set of components comprising various mixers and a frequencydoubler arranged in a particular way. For each radar band signal paththe respective set of components act together, along with the VCO 140,to achieve a composite local oscillator that functions similar to thelinear sweep generator 60 of FIG. 1A. Thus, for each radar band, asF_(LO) is controlled to sweep through its particular range offrequencies at a particular rate, the respective set of components actin concert with one another to function as a composite local oscillatorthat sweeps through a range of frequencies at a particular rate toproduce an intermediate frequency (IF) signal, F_(O), as input to amatched filter.

FIG. 2A depicts major components of the Ka-band signal path from thefront antenna 102 of the radar signal detector of FIG. 1B. The sweepoutput F_(LO) from the VCO 140 is doubled by a frequency doubler 142 andthe second harmonic of this doubled frequency is mixed, at mixer M1F108, with an incoming Ka-band 36 GHz signal. The first intermediatefrequency is in the range of 15.5886 GHz and is mixed, at mixer M3 148,with the doubled F_(LO) to produce a second intermediate frequencysignal in the range of 5.3829 GHz. This second intermediate frequencysignal is mixed, at mixer M4 162, with F_(LO) to produce an outputsignal F_(O) having an intermediate frequency in the range of 280 MHz.This signal is then propagated to the matched filter 170.

As mentioned, the sweep for Ka-band signals may sweep from 33.35 GHz to36.08 GHz. Using the signal path of FIG. 2A, this sweep can beaccomplished when F_(LO) is swept from 4.7242 GHz to 5.1142 GHz.

FIG. 2B depicts major components of the K-band signal path from thefront antenna 102 of the radar signal detector of FIG. 1B. The sweepoutput F_(LO) is doubled by the frequency doubler 142 and its secondharmonic is mixed, at mixer M1F 108, with an incoming K-band 24.25 GHzsignal. The output of the mixer M1F has an intermediate frequency ofabout 5.074 GHz and is mixed, at mixer M4 162, with F_(LO) to produce anoutput signal F_(O) having an intermediate frequency in the range of 280MHz.

As mentioned, the sweep for K-band signals may sweep from 24.035 GHz to24.265 GHz. Using the signal path of FIG. 2B, this sweep can beaccomplished when F_(LO) is swept from 4.751 GHz to 4.797 GHz.

FIG. 2C depicts major components of the Ku-band signal path from thefront antenna 102 of the radar signal detector of FIG. 1B. The sweepoutput F_(LO) is doubled by the frequency doubler 142 and is mixed, atmixer M3 148, with an incoming Ku-band 13.5 GHz signal. The output ofthe mixer M3 has an intermediate frequency of about 4.6866 GHz and ismixed, at mixer M4 162, with F_(LO) to produce an output signal F_(O)having an intermediate frequency in the range of 280 MHz.

As mentioned, the sweep for Ku-band signals may sweep from 13.38 GHz to13.52 GHz. Using the signal path of FIG. 2C, this sweep can beaccomplished when F_(LO) is swept from 4.3667 GHz to 4.4133 GHz.

FIG. 2D depicts major components of the X-band signal path from thefront antenna 102 of the radar signal detector of FIG. 1B. The sweepoutput F_(LO) is additively mixed, at mixer M2 122, with an incomingX-band 10.55 GHz signal to produce a first intermediate frequency signalin the range of 15.685 GHz. The sweep output F_(LO) is also doubled bythe frequency doubler 142 and is mixed, at mixer M3 148, with the firstintermediate frequency signal. The output of the mixer M3 148 has anintermediate frequency of about 5.415 GHz and is mixed, at mixer M4 162,with F_(LO) to produce an output signal F_(O) having an intermediatefrequency in the range of 280 MHz.

As mentioned, the sweep for X-band signals may sweep from 10.49 GHz to10.56 GHz. Using the signal path of FIG. 2D, this sweep can beaccomplished when F_(LO) is swept from 5.105 GHz to 5.14 GHz.

As mentioned above, FIGS. 2A-2D are provided to illustrate an isolatedsignal path for each respective band of radar signals. In these figures,the various switches, amplifiers, and filters have been omitted forclarity and each of the signal paths happen to relate to the frontantenna 102 of FIG. 1B. Similar distinct signal path diagrams can beconstructed as well that relate to the rear antenna 138. To do so, FIGS.2A-2D would be modified to use mixers that are in the signal pathsrelated to the rear antenna 138. For the Ku-band and X-band, nomodifications are needed; the same mixers are used regardless of whetherthe signal path relates to the front antenna 102 or the rear antenna138. For the K-band and Ka-band, the mixer M1F 108 would be replacedwith mixer M1R 130.

The matched filter 170, as mentioned, can be a dispersive delay lineconfigured as a receiver filter. In an embodiment, the filter 170 canhave a matched signal that is an up-chirp with a chirp rate of 3.5MHz/μs. Accordingly, the sweep rate of F_(LO) can be controlled so thatan RF signal at the receiver input is down-converted to the IF andarrives with the matching 3.5 MHz/μs chirp rate. Thus, when the FPGA 176determines the Ka-band using the front antenna 102 is the band ofinterest, it will control all the appropriate switches in FIG. 1B todefine the Ka-band signal path and it will control the VCO 140 toproduce a sweep signal F_(LO) that matches the filter 170.

Using FIG. 2A as an example, the frequency F_(LO) can be characterizedas

$F_{LO} = \frac{{Ka} - F_{O}}{7}$where Ka=36 GHz and F_(O)=280 MHz, for example. Rearranging the termsreveals that F_(O)=Ka−7F_(LO). Thus, any change in the local oscillatorsignal F_(LO) is effectively multiplied by “7” when producing the sweeprate of the composite local oscillator that results in the output chirpF_(O) at the IF frequency of 280 MHz. This can be visualized in FIG. 4Cwhich depicts a graph of the frequency-to-time ratio of the swept localoscillator in accordance with the principles of the present invention.The line 406 represents the sweep local oscillator F_(O) which has aslope, or chirp rate, of Δf/Δt. However, because of the way variousmultiples of F_(LO) are mixed in the signal path of FIG. 2A, theeffective chirp rate 408 of the composite local oscillator comprised ofthe set of components in the signal path of FIG. 2A is 7 times Δf/Δt.When the incoming 36 GHz signal is mixed with the composite localoscillator signal represented by 408 of FIG. 4C, a resulting F_(O)signal is generated as input to the filter 170 and will have a centerfrequency at 280 MHz and be an up-chirp with a chirp rate of 3.5 MHz/μs.Embodiments of the present invention also contemplate slower chirp ratessuch as, for example, 0.15 MHz/μs and 2.85 MHz/μs and even chirp ratesabove 3.5 MHz/μs.

Thus, for the Ka-band the sweep local oscillator signal F_(LO) is notsimply swept at the down chirp rate of −3.5 MHz/μs. Instead, a sweeprate of

$\frac{{- 3.5}\mspace{14mu}{MHz}\text{/}{µs}}{7}$is used to produce a signal F_(O) that is matched to the matching rateof the filter 170.

Accordingly, the F_(LO) signal for the Ka-band is controlled by the FPGA176 to sweep from a starting frequency to a stopping frequency at aspecific sweep rate. The starting and stopping frequencies are selectedto tune through the entire Ka-band and the sweep rate is selected tomatch the impulse response of the filter 170.

As shown in FIGS. 2B-2C, a similar relationship between F_(LO) and areceived radar signal band can be determined for the other three bandssuch as, for example, by:

${{for}\mspace{14mu}{the}\mspace{14mu} K\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{K - F_{O}}{5}$${{for}\mspace{14mu}{the}\mspace{14mu}{Ku}\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{{Ku} - F_{O}}{3}$${{for}\mspace{14mu}{the}\mspace{14mu} X\text{-}{band}\text{:}\mspace{14mu} F_{LO}} = \frac{X - F_{O}}{2}$

This general structure for the F_(LO) signal for each of the radar bandsis the same and depicted in FIG. 5 as the line 506. When the FPGA 176starts the sweep of a particular band, it will control the VCO 140 tostart at a frequency 502, sweep at a particular chirp rate r=Δf/Δt, andstop at a frequency 504. The table below assumes the chirp rate for thecomposite local oscillator to be −3.5 MHz/μs however, as describedabove, the VCO 140 is controlled appropriately to produce a sweepingF_(LO) for each band's signal path in order to achieve this compositechirp rate. The table below provides exemplary values for sweeping thefour radar bands.

VCO CHIRP START RADAR RATE FREQ STOP FREQ SWEEP BAND r f₁ f₂ TIME Ka$\frac{{- 3.5}\mspace{14mu}{MHz}\text{/}{µs}}{7}$ 5.114 GHz 4.724 GHz~780 μs K $\frac{{- 3.5}\mspace{14mu}{MHz}\text{/}{µs}}{5}$ 4.797 GHz4.751 GHz  ~66 μs Ku $\frac{{- 3.5}\mspace{14mu}{MHz}\text{/}{µs}}{3}$4.413 GHz 4.367 GHz  ~40 μs X$\frac{{- 3.5}\mspace{14mu}{MHz}\text{/}{µs}}{2}$ 5.138 GHz 5.103 GHz ~20 μs

Since the sweep signal F_(LO) from the VCO 140 is controlled with highaccuracy, determining a timing of the detected pulse within the sweepduration establishes its frequency. In particular, with respect to FIG.3D, the FPGA 176 determines that a pulse occurs at time T 306 relativeto a sweep starting time of a particular sweep of F_(LO) through one ofthe bands of interest. Such a sweep of F_(LO) is depicted as line 406 inFIG. 4C. In particular, the sweep starts at a time t_(B) 410 and ends ata time t_(E) 412. During that sweep there is a point t_(P) 414 that isthe same distance t_(S) 416 in time from t_(B) 410 as the pulse time T306 is from the sweep start time t_(B) 410.

As shown in the above table, during the sweep of F_(LO), the sweepbegins at a starting frequency f₁ and ends at a stopping frequency f₂.The frequency of F_(LO) at location t_(P) 414 in the sweep is providedby:instanaeous frequency of F _(LO) =f ₁+(t _(S) ×r)This instantaneous value of F_(LO) can then be used to determine thefrequency of the radar signal that corresponds to that portion of thesweep of F_(LO).

For example, if a particular sweep of F_(LO) occurs for the X-band toproduce an output pulse that is 10 μs after the beginning t_(B) 410 ofthe sweep, then t_(S) will equal 10 μs. Using the values f₁=5.114 GHz,

$\left. {r = \frac{{- 3.5}\mspace{14mu}{MHz}\text{/}{µs}}{2}} \right),$and t_(S)=10 μs, will determine that F_(LO)=5.1225 GHz. Using thisinstantaneous frequency for F_(LO), the relationship of FIG. 2D

$\left( {{e.g.},{F_{LO} = \frac{X - F_{O}}{2}}} \right)$and F_(O)=280 MHz, will reveal a value of X=10.525 GHz. Thus, the timingT 306 of the detected output pulse provides an indication that the radarsignal received at an antenna had a frequency of 10.525 GHz.

As mentioned above, there are four different frequency bands that can beswept through in order to detect possible radar gun signals. In additionto these FCC licensed radar bands, it is advantageous to examine othermicrowave frequencies to identify nuisance signals leaking from thelocal oscillators of other radar detectors operating in the vicinity(e.g. 32.55 GHz to 32.65 GHZ and 22.45 GHz to 23.79 GHz). If thesenuisance signals can be identified, the false alarms they generate inKa-band may be suppressed. These problems and techniques for theiramelioration are taught in U.S. Pat. No. 7,579,976, “SYSTEMS AND METHODSFOR DISCRIMINATING SIGNALS IN A MULTI-BAND DETECTOR” which is hereinincorporated by reference in its entirety.

Additionally, in order to determine signal signatures that helpdifferentiate nuisance signals from actual radar gun signals, the fourdifferent bands can be swept through multiple times during one complete“sweep” of the radar detector. Thus, the FPGA 176 can control the VCO140 to sequentially produce appropriate sweep signals (according to theabove table) to sweep through each of the four frequency bands. A sweepthrough all of the four bands can be accomplished in about 1 ms whichallows an opportunity to perform multiple sweeps of the four bandswithin a 16 ms transmission window corresponding to some conventionalbrief-duration radar signal sources.

For example, FIG. 6 depicts a flowchart of an exemplary sweep process.The order in which a front or a rear antenna is selected (e.g., steps602, 612) may vary without departing from the scope of the presentinvention. Similarly, the order in which the four different bands areswept (e.g., steps 604, 606, 608, 610) may vary as well. An examplesweep may start, in step 602, by selecting one of the antennas, such asa front antenna. The VCO 140 can then be controlled to sequentiallysweep through the X-band (step 604), the Ku-band (step 606), the Ka-band(step 608), and the K-band (step 610) as described in more detail above.

Once the four bands have been swept for the selected antenna, then theother antenna, such as the rear antenna, can be selected in step 612.Similar to the sweep using the front antenna, the four bands aresequentially swept in steps 614, 616, 618, and 620. Once the four bandshave been swept for both the front and rear antenna, then adetermination is made in step 622 if the sweeps should be repeated. Forexample, as described earlier, a sweep of all four bands can beaccomplished in about 1 ms and, thus, a sweep for both the front andrear antenna can be accomplished in about 2 ms. If a shortest policeradar transmission signal lasts about 16 ms, then about 6 or 7 sweeps ofall four bands for both antennas can be repeated within that time frame.One of ordinary skill will recognize that the determination in step 622of how many sweeps to repeat can vary without departing from the scopeof the present invention.

Practical design considerations may reduce the number of sweeps that maytheoretically be performed during a predetermined time period (e.g., 16ms) when implementing the sweeping local oscillator. For example, whenchanging from one sweep rate for a particular band to another sweep ratefor another band, the PLL 152 may have a settling time that can beaccounted for. One way to account for the settling time is to start theVCO 140, for a particular band sweep, at a F_(LO) frequency higher thanthe starting frequency f₁ shown in the above table. Following thispractice ensures that the PLL loop has settled and the VCO 140 isaccurately sweeping when the starting frequency f₁ is reached. For anexample explored below, assume the required settling time is 10 μs;during this interval 35 MHz of RF spectrum will be swept.

The DDL SAW filter 170 also has an inherent delay, D_(T), that isaccounted for as well. Assume as an example this delay is 4 μs. Duringthis delay 14 MHz of RF spectrum will be swept (for example, in theKa-band). This delay can be compensated by sweeping 7 MHz before thehighest RF frequency for that band and 7 MHz below the lowest RFfrequency for that band. Further, to accommodate the PLL settling timementioned above, an additional 35 MHz can be incorporated above thenominal start of the band, totaling 42 MHz adjustment of the startingpoint of the RF sweep. Continuing the numerical example, the Ka-band mayhave a high-end RF frequency of 36.08 GHz and a low-end frequency of33.35 GHz. To accommodate the delays described above, the sweepgenerating hardware can be designed to target 36.122 GHz and 33.343 GHzas the start and end points of the RF sweep.

Accounting for the settling time of the PLL 152 and the delay of thefilter 170 plus some “programming overhead” that may be associated withthe time required to load control settings into the reference generator156 may result in a complete sweep of both front and rear antennastaking about 3 ms. Even with this longer sweep period, the detector ofFIG. 1B can complete about 5 of these full sweeps in a 16 ms timeperiod.

Once all the sweeps have been accomplished, then, in step 624, thedetector can provide alarms if a police radar signal was detected andcan also ignore any nuisance signals that, even though detected, do notcorrespond to actual police radar signals.

One known issue with using a mixer in super-heterodyne receivers forcombining a desired RF signal and a local oscillator signal to producean intermediate frequency signal, is that of image signals thatinterfere with receiving and detecting the desired RF signal. Because aradar detector sweeps a spectrum far larger than the final IFcenter-frequency, in the presence of a strong signal it is almostinevitable that the detector must contend with a spurious final IF imageresponse. An example will illustrate: Assume a strong radar signal at35.0 GHz and an equivalent LO at 34.72 GHz, yielding the desiredreceiver response: 35.0−34.72=0.28 GHz. But the detector's LO will alsobe swept through 35.28 GHz, yielding 35.0−35.28=−0.28 GHz, i.e. thereceiver's IF image. Thus, the strong 35 GHz signal will be seen asecond time, but erroneously. Obviously, it is beneficial to design thedetector so that this undesired image signal is attenuated sufficiently.

One of ordinary skill will recognize that there are traditional imagesignal rejection techniques that can be used to filter image signalsprior to the final mixer M4 162 (or more generally, before the mixer 54of FIG. 1A). However, the characteristics of the DDL SAW filter 170provide additional techniques for rejecting image signals.

Assuming, for example, that the DDL filter has delay length of 4 μs anda characteristic chirp rate of 3.5×10¹² Hz/s, then its bandwidth isapproximately (4 μs*3.5 MHz/μs)=14 MHz. If the filter is presented withan up-chirp at the characteristic rate, its resulting output is a narrowpulse lasting a few tens of nanoseconds. Conversely, if presented with adown-chirp, the filter expands rather than compresses the duration ofits output. More specifically, if presented with a 3.5×10¹² Hz/sdown-chirp (e.g. the “image chirp” produced when the receiver sweepsthrough the IF image), the filter output will last about 8 μs, twice thefilter's delay length, and about 28 MHz of spectrum is swept during thisinterval.

The result is that the DDL filter inherently enhances the S/N ratio ofcorrectly-chirped signals but attenuates chirp images by dispersing theenergy, typically yielding more than 20 dB rejection of image chirps.This phenomenon is depicted in FIG. 7 which overlays a strong desiredcompressed chirp 710, having a width T_(cc) 702, with the dispersedresponse 720 arising from reception of an image chirp and having a widthof T_(SIC) 704. It should be noted the receiver and DDL filter arelinear elements so that image suppression prior to the final mixer andchirp-image suppression inherent in the DDL are independent contributorsto image suppression. Additional post-demodulation image attenuation ispossible after the demodulating logarithmic amplifier by virtue of themarked difference in pulse 710 width of the desired response (T_(cc)˜70ns) versus the image pulse width (T_(SIC)˜8 μs). This additionalfiltering may be implemented with analog techniques and/or digitallywithin FPGA algorithms. Each of these image suppression mechanismscombines to improve image signal rejection.

FIG. 8A depicts a flowchart of an exemplary process to implement a fixeddetection window in accordance with the principles of the presentinvention. As mentioned earlier, the output from the filter 170 isreceived by an amplifier 172 that provides input to an analog-to-digitalconverter 174. The FPGA 176 samples the output of the A/D converter 174in time periods that can conveniently be referred to as “buckets” suchthat during a particular bucket, the FPGA 176 determines an input valueof the digital signal provided by the A/D converter 174. Buckets can besequentially numbered so that when the time period for bucket n isfinished, the FPGA 176 can begin sampling the input from the A/Dconverter 174 for bucket n+1. If the time period is known for eachbucket and the bucket number is reset at the start of the sweep, thebucket number can be used to calculate the time since the sweep startusing the formulat _(s) =n×t _(bucket)

where n is the bucket number and t_(bucket) is the period of one bucket.Once the time since the sweep start, t_(s), is known, the techniquesdescribed above can be used to calculate the instantaneous frequency.Therefore, the bucket can be used to determine the detected frequencyduring the detection process.

The overall process of FIG. 8A describes one possible detection routinethat can be performed by the FPGA 176 during a sweep of one of the fourradar frequency bands. In particular the routine for one band can have astarting step 802 along with an initialization of a detected input valuein step 804. Once the routine is started, the input value (to the FPGA176) from the digital output of the A/D converter 174 is determined bythe FPGA 176. More particularly, the FPGA 176, in step 806, calculates adifference between the input value and a noise floor to determine whatis referred to as an “Input Error” value. If the “Input Error” value isgreater than a predetermined threshold, then a desired signal is likelybeing detected and control passes to step 810. If the “Input Error”value does not exceed the predetermined threshold, then control passesto step 808.

In step 808, the FPGA 176 can set a variable “maxLevel” to be the valueof the currently sampled input from the A/D converter 174. Also, avariable “maxBucket” can be set to the value of the current bucketnumber. Control can then loop back to step 806, where the input from theA/D converter 174 is once again analyzed to determine if a detectedsignal is likely present.

Once a likely signal is detected, a timer is started in step 810 and adetermination is made in step 812 whether the value in the currentbucket (i.e., the input from the A/D converter 174) is greater than the“maxLevel”. As the loop of FIG. 8A is repeated, the value of “maxLevel”will be set to the largest value received as input from the A/Dconverter 174 and the value of “maxBucket” will be set to the bucketnumber that corresponds to the largest value received. Thus, if thevalue in the current bucket is larger than the current value of“maxLevel”, the FPGA 176, in step 814, will replace the value for“maxLevel” with the current A/D input value and replace “maxBucket” withthe number of the current bucket.

If the current A/D input is not greater than “maxLevel” or once the“maxLevel” and “maxBucket” values are set, control passes to step 816where the FPGA 176 determines if 1200 ns has elapsed (e.g., a detectionwindow) since the timer was started in step 810. If no, then controlloops back to step 812 to test the current A/D input value. During thetime taken to return to step 812, a new bucket value and bucket numbermay have occurred.

Once the timer has elapsed, according to step 816, the FPGA 176, in step818, places an alert in a queue to possibly be delivered to the userinterface 178. Thus, according to the flowchart of FIG. 8A, the FPGA 176starts a 1200 ns window once the A/D input exceeds a predeterminedthreshold and within that 1200 ns window, the FPGA 176 determines amaximum value that occurred and the time period (i.e., bucket) in whichit occurred. Accordingly, the alert has both a signal strength and atiming tag associated with it.

One of ordinary skill will appreciate that the example 1200 ns detectionwindow can be changed without departing from the scope of the presentinvention and may, for example, vary based on the shape of the sincpulse 304 shown in FIG. 7A. Thus, this detection window can have aduration that captures a desired number of side-lobes 712, 714 andignores the rest. Also, the test in step 816 can be replaced by apredetermined threshold related to a value of the input from the A/Dconverter 174. Thus, rather than having a fixed time duration for adetection window, the FPGA 176 could monitor the signal input from theA/D converter 174 until that input signal drops below a predeterminedthreshold value.

The FPGA 176 repeats the routine of FIG. 8A for each of the four radarbands of interest (and for each of the front and rear antennas). Thus,for each band the FPGA 176 can determine if an alert occurred and alsodetermine for each alert an amplitude of a detected peak signal and whenthat peak signal occurred. As discussed above with respect to FIG. 4C,the time when the peak signal occurred (e.g., time t_(p) 414 of FIG. 4C)reveals the frequency of that peak signal. Accordingly, the FPGA 176 canconstruct, for each sweep of the four radar bands, a snapshot of whatalerts were detected. Because multiple sweeps of the four bands canoccur in a particular time period (e.g., 16 ms), a respective snapshotfor each sweep of the four bands can be constructed each time a sweep ofthe four bands is performed (see FIG. 6, step 622).

The FPGA 176 and/or microprocessor 180 can then compare the differentsnapshots for each of the sweeps to determine which alerts likelycorrespond to police radar signals and which alerts likely correspond tonuisance signals that can be ignored. Nuisance signals can include, forexample, park-assist systems, automatic cruise control radar systems,and harmonics emitted by nearby radar detectors. U.S. Pat. No. 5,852,417describes a variety of nuisance signals and techniques fordiscriminating them from detected signals corresponding to actual policeradar and is incorporated herein by reference.

For example, police radar sources typically emit a signal having a fixedfundamental frequency within their respective band of operation. Whilemany other signal sources emit at a fundamental frequency, they may alsoemit at harmonic frequencies as well. Thus, the FPGA 176 may detect thepresence of a signal at a particular frequency, f_(x), within one of thebands. Using the snapshots of the different sweeps through the bands,the FPGA 176 can also determine if signals at multiples of f_(x) (e.g.,2f_(x), 3f_(x), etc.) were also detected. If signals at harmonicfrequencies of f_(x) were also detected, then the source of the f_(x)signal is likely not a radar gun and an alert at the frequency f_(x) canbe ignored.

Because a snapshot of the approximately 3 GHz of spectrum of the fourradar bands can be generated in about 3 ms, a single snapshot cancapture the occurrence of a nuisance signal and its harmonics even ifthat nuisance signal is short-lived. Also, a nuisance signal and itsaccompanying harmonic signals may typically rise and fall atapproximately the same time and, therefore, it is beneficial to be ableto sweep through a range of frequencies sufficiently fast enough todetect the occurrence of these multiple signals. Thus, in discriminatingsignals to identify alerts which likely correspond to actual policeradar sources, the FPGA 176 can compare data within a snapshot of theswept four radar bands and can compare data within the multiplesnapshots of the four swept radar bands.

Additionally, because it is desirable to detect police radar signals asfar from the police radar gun as possible, the signals of interest maybe subject to significant multipath fading. In such instances, a signalreceived by the detector will vary greatly in intensity over arelatively short time period. Thus, when the detector is tuned toreceive a particular frequency, the received signal at that frequencymay have a negligible amplitude and when the received signal has adetectable amplitude, the detector may no longer be tuned to theappropriate frequency. According to the process described with respectto FIG. 6, the speed at which each sweep of a radar band occurs and therepeated number of sweeps improve the likelihood that even a signalsuffering from multipath fading can be more reliably detected becausethe detector will more likely be tuned to detect the received signalduring one or more periods at which the received signal exhibits adetectable amplitude.

Thus, each respective snapshot can be thought of as a virtual image ofthe signal environment represented by the received electromagneticsignals. Rather than relying on only different sweeps of various bandsof the electromagnetic signal environment that can occur at a rate thatis relatively large as compared to the duration of some of the detectedradar signals, the snapshot, or virtual image, provides informationabout all the swept bands in a timeframe that is beneficial fordetecting short-duration radar signals and/or various nuisance signals.

FIG. 8B depicts a block level diagram of utilizing a dynamicallyadjusted noise floor in accordance with the principles of the presentinvention. In particular, a current value output from the A/D converter174 is provided to a data flip flop (DFF) 830. The output 831 from theDFF 830 is multiplied by 256 and then is provided to an adder 834. Theadder 834 subtracts a noise-related value 835 from the output value ofthe multiplier 832. The output from the adder 834 is passed through aDFF 836 to be one of the input values 846 of a comparator 844. The inputvalue 846 reflects a difference between the output from the multiplier832 and a noise-related value, and can be referred to as “Input Error”as noted above. The noise-related value 835 can be referred to as a“Noise Floor”.

One exemplary noise-related value 835 is shown in FIG. 8B, in which the“Input Error” is first divided by 8, by block 838, and then summed witha previous value 839 of itself using the adder 852. The value 839 can bedivided by 256 by block 842 and may represent a running average of thenoise present in the signal received as input from the A/D converter174.

Returning to the comparator 844, the other input value to the comparator844 is a predetermined threshold value 848. If the “Input Error” 846 isgreater than the predetermined threshold value 848, then a signal foundvalue 852 is output from the comparator 844. This signal found value 852is equivalent to the test analysis of step 806 in FIG. 8A. Thecomponents depicted in FIG. 8B may implement a signal detectiontechnique which relies on a dynamically derived noise floor such that asignal is detected only when an input signal value is greater than thedynamically calculated noise floor by at least a predetermined amount.

In practice, the emitter circuitry of a police radar gun does notinstantly turn on at a desired frequency. Instead, due to thermalexpansion transients, the typical initial emission from the radar gun isat a higher frequency which ramps down quickly to its desired frequencyof operation. This phenomenon can be captured by a radar detectoroperating in accordance with the principles of the present invention.

In particular, if four combined sweeps through four different radarbands constitute a full sweep cycle for an antenna (e.g., a frontantenna), then activation of the emitter circuitry of a radar gun canpotentially be detected in each of the four different sweep cycles. FIG.9 depicts what a detected signal of the activation of a Ka band radargun may look like; it shows detected signals from four different Kasweeps on a single time line. During the first sweep 904, a detectedsignal is determined to occur at t₁ 903 which corresponds to aparticular frequency. Similarly a second signal is detected during thesecond sweep 907 at t₃ 906, a third signal is detected during a thirdsweep 910 at t₅ 909, and a fourth signal is detected during a fourthsweep 913 at t₇ 912.

The FPGA 176 and/or microprocessor 180 can include signal analysisroutines that recognize the pattern of these four signals as potentiallyrepresenting the operation of a police radar gun even though the radargun has yet to emit a radar signal at its desired frequency.

While particular embodiments of the present invention have beenillustrated and described, it would be obvious to those skilled in theart that various other changes and modifications can be made withoutdeparting from the spirit and scope of the invention. It is thereforeintended to cover in the appended claims all such changes andmodifications that are within the scope of this invention.

What is claimed is:
 1. A method of detecting continuous wave policeradar comprising: receiving an input signal from a first antenna, theinput signal comprising a continuous wave emission within at least oneradar band; sweeping a composite local oscillator signal through a rangeof frequencies from a first frequency to a second frequency in apredetermined time period so that the composite local oscillator signalhas a first chirp rate with a first chirp rate magnitude of at least0.15 MHz/μs; mixing the input signal from the first antenna with thesweeping composite local oscillator signal to produce an output signalhaving an intermediate frequency; and determining, by a processorexecuting a signal analysis routine, that the input signal from thefirst antenna includes a police radar signal having a frequency f, basedat least in part on: measuring an amplitude of the output signal for aperiod of time, the period of time comprising a sequential series ofadjacent time periods; determining which particular one of the adjacenttime periods corresponds to a maximum measured amplitude of the outputsignal; determining an amount of time that has elapsed between a startof the sweeping of the composite local oscillator and the particular oneof the adjacent time periods; and calculating the frequency f based onthe amount of time that has elapsed.
 2. The method of claim 1, whereineach of the adjacent time periods comprise a duration d and aresequentially numbered from 1 to n, wherein (n× d)=the period of time theamplitude of the output signal is measured.
 3. The method of claim 2,wherein when the particular one of the adjacent time periods is anm^(th) time period in the sequential series of adjacent time periods,f=m×d.
 4. The method of claim 1, wherein determining that the inputsignal from the first antenna includes a police radar signal having afrequency f further comprises: for each of the adjacent time periods:detecting a time-point when the amplitude of the output signal exceeds apredetermined threshold; monitoring the amplitude of the output signalfor a time-window after the detected time-point; and determining a localmaximum measured output signal value during the time-window; and whereindetermining which particular one of the adjacent time periodscorresponds to a maximum measured amplitude of the output signal isbased on which of the local maximum measured output signal values islargest.
 5. The method of claim 1, further comprising: filtering theoutput signal using a dispersive delay line filter to produce a filteredoutput signal; and determining, by the processor executing the signalanalysis routine, that the input signal from the first antenna includesthe police radar signal having a frequency f, based at least in part on:measuring an amplitude of the filtered output signal for the period oftime, the period of time comprising the sequential series of adjacenttime periods; determining which particular one of the adjacent timeperiods corresponds to a maximum measured amplitude of the filteredoutput signal; determining the amount of time that has elapsed between astart of the sweeping of the composite local oscillator and theparticular one of the adjacent time periods; and calculating thefrequency f based on the amount of time that has elapsed.
 6. The methodof claim 5, wherein determining that the input signal from the firstantenna includes a police radar signal having a frequency f furthercomprises: for each of the adjacent time periods: detecting a time-pointwhen the amplitude of the filtered output signal exceeds a predeterminedthreshold; monitoring the amplitude of the filtered output signal for atime-window after the detected time-point; and determining a localmaximum measured output signal value during the time-window; and whereindetermining which particular one of the adjacent time periodscorresponds to a maximum measured amplitude of the filtered outputsignal is based on which of the local maximum measured output signalvalues is largest.
 7. The method of claim 6, further comprising:converting the filtered output signal from analog to a digital filteredoutput signal.
 8. The method of claim 7, further comprising: an inputerror signal comprising a plurality of sample values, wherein each ofthe plurality of sample values of the input error signal: correspond toan associated sample value of the digital filtered output signal; andits value is equal to a difference between its associated sample valueof the digital filtered output signal and a noise floor amount.
 9. Themethod of claim 8, wherein determining that the input signal from thefirst antenna includes a police radar signal having a frequency ffurther comprises: for each of the adjacent time periods: detecting thetime-point when an input error signal sample value exceeds thepredetermined threshold; monitoring the input error signal for thetime-window after the detected time-point; and determining a localmaximum input error signal sample value during the time-window; andwherein determining which particular one of the adjacent time periodscorresponds to a maximum measured amplitude of the filtered outputsignal is based on which of the local maximum input error signal samplevalues is largest.
 10. The method of claim 1, wherein the first chirprate magnitude is at least 2.85 MHz/μs.
 11. The method of claim 1,wherein the first chirp rate magnitude is at least 3.5 MHz/μs.
 12. Themethod of claim 5, wherein the dispersive delay line filter has a secondchirp rate with a second chirp magnitude equal to the first chirp ratemagnitude and with an inverse slope to that of the first chirp rate. 13.The method of claim 1, wherein the at least one radar band includes fourradar bands, wherein each radar band has a respective starting frequencyand stopping frequency.
 14. A detector for detecting continuous wavepolice radar comprising: a first antenna configured to receive an inputsignal, the input signal comprising a continuous wave emission within atleast one radar band; a composite local oscillator configured to sweep asignal through a range of frequencies from a first frequency to a secondfrequency in a predetermined time period to produce a composite localoscillator signal having a first chirp rate with a first chirp ratemagnitude of at least 0.15 MHz/μs; a mixer configured to mix the inputsignal from the first antenna with the sweeping composite localoscillator signal to produce an output signal having an intermediatefrequency; and a signal analyzer circuit configured to determine whetherthe input signal from the first antenna includes a police radar signalhaving a frequency f, based at least in part on: an amplitude of theoutput signal measured for a period of time, the period of timecomprising a sequential series of adjacent time periods; a determinationof which particular one of the adjacent time periods corresponds to amaximum measured amplitude of the output signal; a determination of anamount of time that has elapsed between a start of the sweeping of thecomposite local oscillator and the particular one of the adjacent timeperiods; and a calculation of the frequency f based on the amount oftime that has elapsed.
 15. The detector of claim 14, wherein each of theadjacent time periods comprise a duration d and are sequentiallynumbered from 1 to n, wherein (n×d)=the period of time the amplitude ofthe output signal is measured.
 16. The detector of claim 15, whereinwhen the particular one of the adjacent time periods is an m^(th) timeperiod in the sequential series of adjacent time periods, f=m×d.
 17. Thedetector of claim 14, wherein the determination that the input signalfrom the first antenna includes the police radar signal having thefrequency f further comprises: for each of the adjacent time periods:detecting a time-point when the amplitude of the output signal exceeds apredetermined threshold; monitoring the amplitude of the output signalfor a time-window after the detected time-point; and determining a localmaximum measured output signal value during the time-window; and whereinthe determination of the particular one of the adjacent time periods isbased on which of the local maximum measured output signal values islargest.
 18. The detector of claim 14, further comprising: a dispersivedelay line filter configured to filter the output signal to produce afiltered output signal; and wherein the signal analyzer is configured todetermine that the input signal from the first antenna includes thepolice radar signal having the frequency f, based at least in part on:measuring an amplitude of the filtered output signal for the period oftime, the period of time comprising the sequential series of adjacenttime periods; determining which particular one of the adjacent timeperiods corresponds to a maximum measured amplitude of the filteredoutput signal; determining the amount of time that has elapsed between astart of the sweeping of the composite local oscillator and theparticular one of the adjacent time periods; and calculating thefrequency f based on the amount of time that has elapsed.
 19. Thedetector of claim 18, wherein determining that the input signal from thefirst antenna includes the police radar signal having the frequency ffurther comprises: for each of the adjacent time periods: detecting atime-point when the amplitude of the filtered output signal exceeds apredetermined threshold; monitoring the amplitude of the filtered outputsignal for a time-window after the detected time-point; and determininga local maximum measured output signal value during the time-window; andwherein determining which particular one of the adjacent time periodscorresponds to a maximum measured amplitude of the filtered outputsignal is based on which of the local maximum measured output signalvalues is largest.
 20. The detector of claim 19, further comprising: ananalog-to-digital converter configured to convert the filtered outputsignal from analog to a digital filtered output signal.
 21. The detectorof claim 20, further comprising: an input error signal comprising aplurality of sample values, wherein each of the plurality of samplevalues of the input error signal: corresponds to an associated samplevalue of the digital filtered output signal; and its value is equal to adifference between its associated sample value of the digital filteredoutput signal and a noise floor amount.
 22. The detector of claim 21,wherein the determination that the input signal from the first antennaincludes the police radar signal having the frequency f furthercomprises: for each of the adjacent time periods: detecting thetime-point when an input error signal sample value exceeds thepredetermined threshold; monitoring the input error signal for thetime-window after the detected time-point; and determining a localmaximum input error signal sample value during the time-window; andwherein determining the particular one of the adjacent time periodscorresponds to a maximum measured amplitude of the filtered outputsignal is based on which of the local maximum input error signal samplevalues is largest.
 23. The detector of claim 14, wherein the first chirprate magnitude is at least 2.85 MHz/μs.
 24. The detector of claim 14,wherein the first chirp rate magnitude is at least 3.5 MHz/μs.
 25. Thedetector of claim 19, wherein the dispersive delay line filter has asecond chirp rate with a second chirp magnitude equal to the first chirprate magnitude and with an inverse slope to that of the first chirprate.
 26. The detector of claim 14, wherein the at least one radar bandincludes four radar bands, wherein each radar band has a respectivestarting frequency and stopping frequency.